1. Field of the Invention
The present invention relates generally to ultra-wideband communication systems, and, in particular, to a receiver for use in an ultra-wideband communication system adapted to determine the angle of arrival of an RF signal.
2. Description of the Related Art
In general, in the descriptions that follow, we will italicize the first occurrence of each special term of art which should be familiar to those skilled in the art of ultra-wideband (“UWB”) communication systems. In addition, when we first introduce a term that we believe to be new or that we will use in a context that we believe to be new, we will bold the term and provide the definition that we intend to apply to that term. In addition, throughout this description, we will sometimes use the terms assert and negate when referring to the rendering of a signal, signal flag, status bit, or similar apparatus into its logically true or logically false state, respectively, and the term toggle to indicate the logical inversion of a signal from one logical state to the other. Alternatively, we may refer to the mutually exclusive boolean states as logic_0 and logic_1. Of course, as is well known, consistent system operation can be obtained by reversing the logic sense of all such signals, such that signals described herein as logically true become logically false and vice versa. Furthermore, it is of no relevance in such systems which specific voltage levels are selected to represent each of the logic states.
In general, in an ultra-wideband (“UWB”) communication system, a series of special processing steps are performed by a UWB transmitter to prepare payload data for transmission via a packet-based UWB channel. Upon reception, a corresponding series of reversing steps are performed by a UWB receiver to recover the data payload. Details of both series of processing steps are fully described in IEEE Standards 802.15.4 (“802.15.4”) and 802.15.4a (“802.15.4a”), copies of which are submitted herewith and which are expressly incorporated herein in their entirety by reference. As is known, these Standards describe required functions of both the transmit and receive portions of the system, but specify implementation details only of the transmit portion of the system, leaving to implementers the choice of how to implement the receive portion.
One or more of us have developed certain improvements for use in UWB communication systems, which improvements are fully described in the following pending applications or issued patents, all of which are expressly incorporated herein in their entirety:
“A Method and Apparatus for Generating Codewords”, U.S. Pat. No. 7,787,544, issued 31 Jul. 2010;
“A Method and Apparatus for Generating Codewords”, application Ser. No. 11/309,222, filed 13 Jul. 2006, now abandoned;
“A Method and Apparatus for Transmitting and Receiving Convolutionally Coded Data”, U.S. Pat. No. 7,636,397, issued 22 Dec. 2009;
“A Method and Apparatus for Transmitting and Receiving Convolutionally Coded Data”, U.S. Pat. No. 8,358,709, issued 22 Jan. 2013; and
“Convolution Code for Use in a Communication System”, U.S. Pat. No. 8,677,224, issued 18 Mar. 2014.
One particular problem in multi-path, spread-spectrum systems, including UWB, is channel-induced noise present in the received signal. One common technique for significantly reducing the noise level relative to the receive level is to develop, during reception of a training sequence portion of the preamble of each transmitted packet, an estimate of the channel impulse response (“CIR”). Following detection in the received packet of the start-of-frame delimiter (“SFD”), the best CIR estimate is reversed in time and the complex conjugate is developed. This conjugate CIR estimate is thereafter convolved with the payload portion of the packet using a channel matched filter (“CMF”). Shown in FIG. 1 is a UWB receiver 10 adapted to operate in this manner. As is known, the signal received via an antenna 12 is continuously conditioned by a filter 14. During reception of the training sequence, channel estimator 16 develops from the conditioned signal the conjugate CIR estimate. During reception of the payload data, detector 18 employs a CMF (not shown) to convolve the conditioned signal with the conjugate CIR estimate, thereby significantly improving the signal-to-noise ratio (“SNR”) and facilitating recovery of the payload data. See, also, “Efficient Back-End Channel Matched Filter (CMF)”, U.S. Pat. No. 7,349,461, issued 25 Mar. 2008.
As noted in 802.15.4a, § 5.5.7.1, “UWB devices that have implemented optional ranging support are called ranging-capable devices (“RDEVs”).” (Emphasis in original.) For certain applications, such RDEVs are commonly implemented in the form of a relatively compact, autonomous radio-frequency identification (“RFID”) tag or the like. Due to the small form factor and limited power supply, it is especially important to select circuit implementations that provide maximum performance at minimum power. Unfortunately, in known implementations of the UWB receiver, improvements in performance usually come at the expense of power. For example, it is known that a rake filter provides good performance in multi-path, spread-spectrum systems such as UWB. See, e.g., slide 21 of “The ParthusCeva Ultra Wideband PHY Proposal”, IEEE P802.15 Working Group for Wireless Personal Area Networks, March 2003, a copy of which is submitted wherewith and which is expressly incorporated herein in its entirety by reference. However, known rake filter implementations tend to consume significantly more power than other prior art techniques.
In ranging systems, as in other RF systems, the receiver must coordinate its internal operation to the signal being received from the transmitter. In general, the receiver must achieve synchronism with the received carrier signal, a process referred to as carrier recovery. In addition, the receiver must further achieve synchronism with the information signals superimposed on the carrier, a process referred to as timing recovery. We submit that prior art techniques for performing both carrier recovery and timing recovery in the digital domain are less than optimum.
In the RF system topology shown in FIG. 15, it can be seen that, because of the non-zero angle of incidence, θ, the RF signal will arrive at one antenna before the other. In particular, it can be seen that the path to antenna A is greater than to antenna B by p=d*sin(θ). In order to calculate θ, the angle of incidence, the time difference of arrival could be found. If d is relatively large then this would provide quite an accurate estimate of θ. On the other hand, if d is small the estimate turns out to be highly error prone.
FIG. 16 shows two receivers, 70a and 70b, which are clocked from the same crystal 72. If the same crystal 72 clocks identical phase locked loops (“PLLs”), 74a and 74b, the generated carriers that are supplied to the respective down converter mixers, 76ac-76as and 76bc-76as, will have the same phase. The RF signal will arrive at a slightly later time at antenna A than antenna B, so it will encounter a down converter carrier phase that is different in each of the mixers 76. If the baseband processors, 78a and 78b, are capable of calculating the complex impulse response of the channel, that impulse response will have a different in-phase (“I”) to quadrature (“Q”) ratio I/Q which is equal to the phase delay caused by the signal travelling the extra distance, p, before encountering the mixer 70a and being down-converted by the carrier. If the carrier frequency is high, e.g., 4 GHz or 6.5 GHz, then quite small distances, p, will lead to a relatively large carrier phase difference.
                              sin          ⁢                                          ⁢          θ                =                  p          d                                    [                  Eq          .                                          ⁢          1                ]                                λ        =                  c          f                                    [                  Eq          .                                          ⁢          2                ]            
where:                f is the carrier frequency,        c is the speed of light, and        λ is the carrier wavelength.        
                              α                      2            ⁢            π                          =                  p          λ                                    [                  Eq          .                                          ⁢          3                ]            
where:                α is the phase difference between the two carriers for the same point on the incident RF signal.        
                                                        p              =                              αλ                                  2                  ⁢                  π                                                                                        (                              from                ⁢                                                                  ⁢                                  Eq                  .                                                                          ⁢                  2                                ⁢                                                                  ⁢                and                ⁢                                                                  ⁢                                  Eq                  .                                                                          ⁢                  3                                            )                                                          [                  Eq          .                                          ⁢          4                ]                                                                                    sin                ⁢                                                                  ⁢                θ                            =                              αλ                                  2                  ⁢                  π                  ⁢                                                                          ⁢                  d                                                                                        (                              from                ⁢                                                                  ⁢                                  Eq                  .                                                                          ⁢                  1                                ⁢                                                                  ⁢                and                ⁢                                                                  ⁢                                  Eq                  .                                                                          ⁢                  4                                            )                                                          [                  Eq          .                                          ⁢          5                ]                                                                    θ              =                                                sin                                      -                    1                                                  ⁢                                  αλ                                      2                    ⁢                    π                    ⁢                                                                                  ⁢                    d                                                                                                          (                              from                ⁢                                                                  ⁢                                  Eq                  .                                                                          ⁢                  5                                            )                                                          [                  Eq          .                                          ⁢          6                ]            
If, in Eq. 6, d is set to be a half wavelength, then FIG. 17 shows the relationship between α, the phase difference of the impulse responses, and θ, the angle of incidence. Note that the slope of the dark grey section is approximately 3, whereas the slope of the lighter grey section is 0.6, i.e., 5 times worse. If, however, d is set to be a one wavelength, then FIG. 18 shows the relationship between α and θ. Note that, at this separation, there is an ambiguity in that each phase relationship has two possible angles of incidence. As can be seen from FIG. 19, as the antennae are moved further apart, say to 3 wavelengths, the ambiguity only increases.
We submit that the larger separation of one wavelength or more is advantageous for two reasons: first, the slope of the angle of incidence curve versus phase change curve is larger and stays larger for longer, thereby allowing more accurate determination of angle of incidence; and second, as the antennas get closer together, their near fields interfere and their performance starts to affect each other. This is particularly the case when the separation is lower than one wavelength.
We submit that what is needed is an improved method and apparatus for use in the receiver of a UWB communication system to determine angle of incidence. In particular, we submit that such a method and apparatus should provide performance generally comparable to the best prior art techniques but more efficiently than known implementations of such prior art techniques.